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AD636 데이터시트 PDF




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부품번호 AD636 기능
기능 Low Level/ True RMS-to-DC Converter
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AD636 데이터시트, 핀배열, 회로
a
Low Level,
True RMS-to-DC Converter
AD636
FEATURES
True RMS-to-DC Conversion
200 mV Full Scale
Laser-Trimmed to High Accuracy
0.5% Max Error (AD636K)
1.0% Max Error (AD636J)
Wide Response Capability:
Computes RMS of AC and DC Signals
1 MHz –3 dB Bandwidth: V RMS >100 mV
Signal Crest Factor of 6 for 0.5% Error
dB Output with 50 dB Range
Low Power: 800 A Quiescent Current
Single or Dual Supply Operation
Monolithic Integrated Circuit
Low Cost
Available in Chip Form
PIN CONNECTIONS &
FUNCTIONAL BLOCK DIAGRAM
IOUT
VIN 1
ABSOLUTE
VALUE
NC 2 AD636
–VS 3
CAV 4
SQUARER
DIVIDER
dB 5
CURRENT
MIRROR
BUF OUT 6
BUF IN 7
+
BUF
10k
10k
14 +VS
13 NC
12 NC
11 NC
COMMON
RL BUF IN
10k
AD636
+–
BUF
BUF OUT
CURRENT
MIRROR
10k
10 COMMON
9 RL
8 IOUT
+VS
VIN
SQUARER
DIVIDER
ABSOLUTE
VALUE
dB
CAV
NC = NO CONNECT
–VS
PRODUCT DESCRIPTION
The AD636 is a low power monolithic IC which performs true
rms-to-dc conversion on low level signals. It offers performance
which is comparable or superior to that of hybrid and modular
converters costing much more. The AD636 is specified for a
signal range of 0 mV to 200 mV rms. Crest factors up to 6 can
be accommodated with less than 0.5% additional error, allowing
accurate measurement of complex input waveforms.
The low power supply current requirement of the AD636, typi-
cally 800 µA, allows it to be used in battery-powered portable
instruments. A wide range of power supplies can be used, from
± 2.5 V to ±16.5 V or a single +5 V to +24 V supply. The input
and output terminals are fully protected; the input signal can
exceed the power supply with no damage to the device (allowing
the presence of input signals in the absence of supply voltage)
and the output buffer amplifier is short-circuit protected.
The AD636 includes an auxiliary dB output. This signal is
derived from an internal circuit point which represents the loga-
rithm of the rms output. The 0 dB reference level is set by an
externally supplied current and can be selected by the user
to correspond to any input level from 0 dBm (774.6 mV) to
–20 dBm (77.46 mV). Frequency response ranges from 1.2 MHz
at a 0 dBm level to over 10 kHz at –50 dBm.
The AD636 is designed for ease of use. The device is factory-
trimmed at the wafer level for input and output offset, positive
and negative waveform symmetry (dc reversal error), and full-
scale accuracy at 200 mV rms. Thus no external trims are re-
quired to achieve full-rated accuracy.
AD636 is available in two accuracy grades; the AD636J total
error of ± 0.5 mV ± 0.06% of reading, and the AD636K
is accurate within ± 0.2 mV to ± 0.3% of reading. Both versions
are specified for the 0°C to +70°C temperature range, and are
offered in either a hermetically sealed 14-pin DIP or a 10-lead
TO-100 metal can. Chips are also available.
PRODUCT HIGHLIGHTS
1. The AD636 computes the true root-mean-square of a com-
plex ac (or ac plus dc) input signal and gives an equivalent dc
output level. The true rms value of a waveform is a more
useful quantity than the average rectified value since it is a
measure of the power in the signal. The rms value of an
ac-coupled signal is also its standard deviation.
2. The 200 millivolt full-scale range of the AD636 is compatible
with many popular display-oriented analog-to-digital con-
verters. The low power supply current requirement permits
use in battery powered hand-held instruments.
3. The only external component required to perform measure-
ments to the fully specified accuracy is the averaging capaci-
tor. The value of this capacitor can be selected for the desired
trade-off of low frequency accuracy, ripple, and settling time.
4. The on-chip buffer amplifier can be used to buffer either the
input or the output. Used as an input buffer, it provides
accurate performance from standard 10 Minput attenua-
tors. As an output buffer, it can supply up to 5 milliamps of
output current.
5. The AD636 will operate over a wide range of power supply
voltages, including single +5 V to +24 V or split ± 2.5 V to
± 16.5 V sources. A standard 9 V battery will provide several
hundred hours of continuous operation.
REV. B
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700 World Wide Web Site: http://www.analog.com
Fax: 781/326-8703
© Analog Devices, Inc., 1999




AD636 pdf, 반도체, 판매, 대치품
AD636
APPLYING THE AD636
The input and output signal ranges are a function of the supply
voltages as detailed in the specifications. The AD636 can also
be used in an unbuffered voltage output mode by disconnecting
the input to the buffer. The output then appears unbuffered
across the 10 kresistor. The buffer amplifier can then be used
for other purposes. Further, the AD636 can be used in a current
output mode by disconnecting the 10 kresistor from the
ground. The output current is available at Pin 8 (Pin 10 on the
“H” package) with a nominal scale of 100 µA per volt rms input,
positive out.
OPTIONAL TRIMS FOR HIGH ACCURACY
If it is desired to improve the accuracy of the AD636, the exter-
nal trims shown in Figure 2 can be added. R4 is used to trim the
offset. The scale factor is trimmed by using R1 as shown. The
insertion of R2 allows R1 to either increase or decrease the scale
factor by ± 1.5%.
The trimming procedure is as follows:
1. Ground the input signal, VIN, and adjust R4 to give zero
volts output from Pin 6. Alternatively, R4 can be adjusted to
give the correct output with the lowest expected value of VIN.
2. Connect the desired full-scale input level to VIN, either dc or
a calibrated ac signal (1 kHz is the optimum frequency);
then trim R1 to give the correct output from Pin 6, i.e.,
200 mV dc input should give 200 mV dc output. Of course,
a ± 200 mV peak-to-peak sine wave should give a 141.4 mV
dc output. The remaining errors, as given in the specifica-
tions, are due to the nonlinearity.
SCALE
FACTOR
ADJUST
VIN
R1
200
؎1.5%
–VS
CAV
–+
1 ABSOLUTE
VALUE
2 AD636
3 SQUARER
DIVIDER
4
14
13
12
11
+VS
VOUT
5
CURRENT
10
MIRROR
R2
154
+VS
6+
9
10k
7 BUF
8
10k
R3
470k
R4
500k
–VS
OFFSET
ADJUST
Figure 2. Optional External Gain and Output Offset Trims
SINGLE SUPPLY CONNECTION
The applications in Figures 1 and 2 assume the use of dual
power supplies. The AD636 can also be used with only a single
positive supply down to +5 volts, as shown in Figure 3. Figure 3
is optimized for use with a 9 volt battery. The major limitation
of this connection is that only ac signals can be measured since
the input stage must be biased off ground for proper operation.
This biasing is done at Pin 10; thus it is critical that no extrane-
ous signals be coupled into this point. Biasing can be accom-
plished by using a resistive divider between +VS and ground.
The values of the resistors can be increased in the interest of
lowered power consumption, since only 1 microamp of current
flows into Pin 10 (Pin 2 on the “H” package). Alternately, the
COM pin of some CMOS ADCs provides a suitable artificial
ground for the AD636. AC input coupling requires only capaci-
tor C2 as shown; a dc return is not necessary as it is provided
internally. C2 is selected for the proper low frequency break
point with the input resistance of 6.7 k; for a cut-off at 10 Hz,
C2 should be 3.3 µF. The signal ranges in this connection are
slightly more restricted than in the dual supply connection. The
load resistor, RL, is necessary to provide current sinking capability.
C2
3.3F
VIN
NONPOLARIZED
VOUT
RL
10kto 1k
CAV
–+
1 ABSOLUTE
VALUE
2 AD636
3 SQUARER
DIVIDER
4
14
13
12
11
5
CURRENT
10
MIRROR
6
+
9
10k
7 BUF
8
10k
+VS
0.1F
20k
0.1F
39k
Figure 3. Single Supply Connection
CHOOSING THE AVERAGING TIME CONSTANT
The AD636 will compute the rms of both ac and dc signals. If
the input is a slowly-varying dc voltage, the output of the AD636
will track the input exactly. At higher frequencies, the average
output of the AD636 will approach the rms value of the input
signal. The actual output of the AD636 will differ from the ideal
output by a dc (or average) error and some amount of ripple, as
demonstrated in Figure 4.
EO
IDEAL
EO DC ERROR = EO – EO (IDEAL)
DOUBLE-FREQUENCY AVERAGE EO = EO
RIPPLE
TIME
Figure 4. Typical Output Waveform for Sinusoidal Input
The dc error is dependent on the input signal frequency and the
value of CAV. Figure 5 can be used to determine the minimum
value of CAV which will yield a given % dc error above a given
frequency using the standard rms connection.
The ac component of the output signal is the ripple. There are
two ways to reduce the ripple. The first method involves using
a large value of CAV. Since the ripple is inversely proportional
to CAV, a tenfold increase in this capacitance will effect a tenfold
reduction in ripple. When measuring waveforms with high crest
factors, (such as low duty cycle pulse trains), the averaging time
constant should be at least ten times the signal period. For
example, a 100 Hz pulse rate requires a 100 ms time constant,
which corresponds to a 4 µF capacitor (time constant = 25 ms
per µF).
–4– REV. B

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AD636 전자부품, 판매, 대치품
AD636
resemble low duty cycle pulse trains, such as those occurring in
switching power supplies and SCR circuits, have high crest
factors. For example, a rectangular pulse train with a 1% duty
cycle has a crest factor of 10 (C.F. = 1 η ).
Circuit Description
The input voltage, VIN, is ac coupled by C4 while resistor R8,
together with diodes D1, and D2, provide high input voltage
protection.
Figure 13 is a curve of reading error for the AD636 for a 200 mV
rms input signal with crest factors from 1 to 7. A rectangular
pulse train (pulse width 200 µs) was used for this test since it is
the worst-case waveform for rms measurement (all the energy is
contained in the peaks). The duty cycle and peak amplitude
were varied to produce crest factors from 1 to 7 while maintain-
ing a constant 200 mV rms input amplitude.
0.5
VP
0
T
0 200s EO
200s
= DUTY CYCLE =
T
CF = 1/
EIN (rms) = 200mV
–0.5
The buffer’s output, Pin 6, is ac coupled to the rms converter’s
input (Pin 1) by capacitor C2. Resistor, R9, is connected between
the buffer’s output, a Class A output stage, and the negative output
swing. Resistor R1, is the amplifier’s “bootstrapping” resistor.
With this circuit, single supply operation is made possible by
setting “ground” at a point between the positive and negative
sides of the battery. This is accomplished by sending 250 µA
from the positive battery terminal through resistor R2, then
through the 1.2 volt AD589 bandgap reference, and finally back
to the negative side of the battery via resistor R10. This sets
ground at 1.2 volts +3.18 volts (250 µA × 12.7 k) = 4.4 volts
below the positive battery terminal and 5.0 volts (250 µA × 20 k)
above the negative battery terminal. Bypass capacitors C3 and
C5 keep both sides of the battery at a low ac impedance to
ground. The AD589 bandgap reference establishes the 1.2 volt
regulated reference voltage which together with resistor R3 and
trimming potentiometer R4 set the zero dB reference current IREF.
–1.0
1
23456
CREST FACTOR
Figure 13. Error vs. Crest Factor
7
A COMPLETE AC DIGITAL VOLTMETER
Figure 14 shows a design for a complete low power ac digital
voltmeter circuit based on the AD636. The 10 Minput
attenuator allows full-scale ranges of 200 mV, 2 V, 20 V and
200 V rms. Signals are capacitively coupled to the AD636 buffer
amplifier, which is connected in an ac bootstrapped configura-
tion to minimize loading. The buffer then drives the 6.7 k
input impedance of the AD636. The COM terminal of the ADC
chip provides the false ground required by the AD636 for single
supply operation. An AD589 1.2 volt reference diode is used to
provide a stable 100 millivolt reference for the ADC in the lin-
ear rms mode; in the dB mode, a 1N4148 diode is inserted in
series to provide correction for the temperature coefficient of the
dB scale factor. Calibration of the meter is done by first adjust-
ing offset pot R17 for a proper zero reading, then adjusting the
R13 for an accurate readout at full scale.
Calibration of the dB range is accomplished by adjusting R9 for
the desired 0 dB reference point, then adjusting R14 for the
desired dB scale factor (a scale of 10 counts per dB is convenient).
Total power supply current for this circuit is typically 2.8 mA
using a 7106-type ADC.
A LOW POWER, HIGH INPUT IMPEDANCE dB METER
Introduction
The portable dB meter circuit featured here combines the func-
tions of the AD636 rms converter, the AD589 voltage reference,
and a µA776 low power operational amplifier. This meter offers
excellent bandwidth and superior high and low level accuracy
while consuming minimal power from a standard 9 volt transis-
tor radio battery.
In this circuit, the built-in buffer amplifier of the AD636 is used
as a “bootstrapped” input stage increasing the normal 6.7 k
input Z to an input impedance of approximately 1010 .
Performance Data
0 dB Reference Range = 0 dBm (770 mV) to –20 dBm
(77 mV) rms
0 dBm = 1 milliwatt in 600
Input Range (at IREF = 770 mV) = 50 dBm
Input Impedance = approximately 1010
VSUPPLY Operating Range +5 V dc to +20 V dc
IQUIESCENT = 1. 8 mA typical
Accuracy with 1 kHz sine wave and 9 volt dc supply:
0 dB to –40 dBm ± 0.1 dBm
0 dBm to –50 dBm ± 0.15 dBm
+10 dBm to –50 dBm ± 0.5 dBm
Frequency Response ؎3 dBm
Input
0 dBm = 5 Hz to 380 kHz
–10 dBm = 5 Hz to 370 kHz
–20 dBm = 5 Hz to 240 kHz
–30 dBm = 5 Hz to 100 kHz
–40 dBm = 5 Hz to 45 kHz
–50 dBm = 5 Hz to 17 kHz
Calibration
1. First calibrate the zero dB reference level by applying a 1 kHz
sine wave from an audio oscillator at the desired zero dB
amplitude. This may be anywhere from zero dBm (770 mV
rms – 2.2 volts p-p) to –20 dBm (77 mV rms 220 mV – p-p).
Adjust the IREF cal trimmer for a zero indication on the analog
meter.
2. The final step is to calibrate the meter scale factor or gain.
Apply an input signal –40 dB below the set zero dB reference
and adjust the scale factor calibration trimmer for a 40 µA
reading on the analog meter.
The temperature compensation resistors for this circuit may be
purchased from: Tel Labs Inc., 154 Harvey Road, P.O. Box 375,
Londonderry, NH 03053, Part #Q332A 2 k1% +3500 ppm/°C
or from Precision Resistor Company, 109 U.S. Highway 22, Hill-
side, NJ 07205, Part #PT146 2 k1% +3500 ppm/°C.
REV. B
–7–

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