PDF AD693 Data sheet ( Hoja de datos )

Número de pieza	AD693
Descripción	Loop-Powered 4.20 mA Sensor Transmitter
Fabricantes	Analog Devices
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No Preview Available ! a FEATURES Instrumentation Amplifier Front End Loop-Powered Operation Precalibrated 30 mV or 60 mV Input Spans Independently Adjustable Output Span and Zero Precalibrated Output Spans: 4–20 mA Unipolar 0–20 mA Unipolar 12 ؎ 8 mA Bipolar Precalibrated 100 ⍀ RTD Interface 6.2 V Reference with Up to 3.5 mA of Current Available Uncommitted Auxiliary Amp for Extra Flexibility Optional External Pass Transistor to Reduce Self-Heating Errors Loop-Powered 4–20 mA Sensor Transmitter AD693 FUNCTIONAL BLOCK DIAGRAM PRODUCT DESCRIPTION The AD693 is a monolithic signal conditioning circuit which accepts low-level inputs from a variety of transducers to control a standard 4–20 mA, two-wire current loop. An on-chip voltage reference and auxiliary amplifier are provided for transducer excitation; up to 3.5 mA of excitation current is available when the device is operated in the loop-powered mode. Alternatively, the device may be locally powered for three-wire applications when 0–20 mA operation is desired. Precalibrated 30 mV and 60 mV input spans may be set by simple pin strapping. Other spans from 1 mV to 100 mV may be realized with the addition of external resistors. The auxiliary amplifier may be used in combination with on-chip voltages to provide six precalibrated ranges for 100 Ω RTDs. Output span and zero are also determined by pin strapping to obtain the standard ranges: 4–20mA, 12 ± 8 mA and 0–20 mA. Active laser trimming of the AD693’s thin-film resistors result in high levels of accuracy without the need for additional adjustments and calibration. Total unadjusted error is tested on every device to be less than 0.5% of full scale at +25°C, and less than 0.75% over the industrial temperature range. Residual nonlinearity is under 0.05%. The AD693 also allows for the use of an external pass transistor to further reduce errors caused by self-heating. For transmission of low-level signals from RTDs, bridges and pressure transducers, the AD693 offers a cost-effective signal conditioning solution. It is recommended as a replacement for discrete designs in a variety of applications in process control, factory automation and system monitoring. The AD693 is packaged in a 20-pin ceramic side-brazed DIP, 20-pin Cerdip, and 20-pin LCCC and is specified over the –40°C to +85°C industrial temperature range. REV. A Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. PRODUCT HIGHLIGHTS 1. The AD693 is a complete monolithic low-level voltage-to- current loop signal conditioner. 2. Precalibrated output zero and span options include 4–20 mA, 0–20 mA, and 12 ± 8 mA in two- and three-wire configurations. 3. Simple resistor programming adds a continuum of ranges to the basic 30 mV and 60 mV input spans. 4. The common-mode range of the signal amplifier input extends from ground to near the device’s operating voltage. 5. Provision for transducer excitation includes a 6.2 V reference output and an auxiliary amplifier which may be configured for voltage or current output and signal amplification. 6. The circuit configuration permits simple linearization of bridge, RTD, and other transducer signals. 7. A monitored output is provided to drive an external pass transistor. This feature off-loads power dissipation to extend the temperature range of operation, enhance reliability, and minimize self-heating errors. 8. Laser-wafer trimming results in low unadjusted errors and affords precalibrated input and output spans. 9. Zero and span are independently adjustable and noninteractive to accommodate transducers or user defined ranges. 10. Six precalibrated temperature ranges are available with a 100 Ω RTD via pin strapping. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 617/329-4700 Fax: 617/326-8703 1 page AD693 FUNCTIONAL DESCRIPTION The operation of the AD693 can be understood by dividing the circuit into three functional parts (see Figure 9). First, an instrumentation amplifier front-end buffers and scales the low- level input signal. This amplifier drives the second section, a V/I converter, which provides the 4-to-20mA loop current. The third section, a voltage reference and resistance divider, provides application voltages for setting the various “live zero” currents. In addition to these three main sections, there is an on-chip auxiliary amplifier which can be used for transducer excitation. VOLTAGE-TO-CURRENT (V/I) CONVERTER The output NPN transistor for the V/I section sinks loop current when driven on by a high gain amplifier at its base. The input for this amplifier is derived from the difference in the outputs of the matched preamplifiers having gains, G2. This difference is caused to be small by the large gain, +A, and the negative feedback through the NPN transistor and the loop current sampling resistor between IIN and Boost. The signal across this resistor is compared to the input of the left preamp and servos the loop current until both signals are equal. Accurate voltage-to-current transformation is thereby assured. The preamplifiers employ a special design which allows the active feedback amplifier to operate from the most positive point in the circuit, IIN. The V/I stage is designed to have a nominal transconductance of 0.2666 A/V. Thus, a 75 mV signal applied to the inputs of the V/I (Pin 16, noninverting; Pin 12, inverting) results in a full-scale output current of 20 mA. The current limiter operates as follows: the output of the feed- back preamp is an accurate indication of the loop current. This output is compared to an internal setpoint which backs off the drive to the NPN transistor when the loop current approaches 25 mA. As a result, the loop and the AD693 are protected from the consequences of voltage overdrive at the V/I input. VOLTAGE REFERENCE AND DIVIDER A stabilized bandgap voltage reference and laser-trimmed resistor divider provide for both transducer excitation as well as precalibrated offsets for the V/I converter. When not used for external excitation, the reference should be loaded by approxi- mately 1 mA (6.2 kΩ to common). The 4 mA and 12 mA taps on the resistor divider correspond to –15 mV and –45 mV, respectively, and result in a live zero of 4 mA or 12 mA of loop current when connected to the V/I converter’s inverting input (Pin 12). Arranging the zero offset in this way makes the zero signal output current independent of input span. When the input to the signal amp is zero, the noninverting input of the V/I is at 6.2 V. Since the standard offsets are laser trimmed at the factory, adjustment is seldom necessary except to accommodate the zero offset of the actual source. (See “Adjusting Zero.”) SIGNAL AMPLIFIER The Signal Amplifier is an instrumentation amplifier used to buffer and scale the input to match the desired span. Inputs applied to the Signal Amplifier (at Pins 17 and 18) are amplified and referred to the 6.2 V reference output in much the same way as the level translation occurs in the V/I converter. Signals from the two preamplifiers are subtracted, the difference is amplified, and the result is fed back to the upper preamp to minimize the difference. Since the two preamps are identical, this minimum will occur when the voltage at the upper preamp just matches the differential input applied to the Signal Amplifier at the left. Since the signal which is applied to the V/I is attenuated across the two 800 Ω resistors before driving the upper preamp, it will necessarily be an amplified version of the signal applied between Pins 17 and 18. By changing this attenuation, you can control the span referred to the Signal Amplifier. To illustrate: a 75 mV signal applied to the V/I results in a 20 mA loop current. Nominally, 15 mV is applied to offset the zero to 4 mA leaving a 60 mV range to correspond to the span. And, since the nominal attenuation of the resistors connected to Pins 16, 15 and 14 is 2.00, a 30 mV input signal will be doubled to result in 20 mA of loop current. Shorting Pins 15 and 16 results in unity gain and permits a 60 mV input span. Other choices of span may be implemented with user supplied resistors to modify the attenuation. (See section “Adjusting Input Span.”) The Signal Amplifier is specially designed to accommodate a large common-mode range. Common-mode signals anywhere up to and beyond the 6.2 V reference are easily handled as long as VIN is sufficiently positive. The Signal Amplifier is biased with respect to VIN and requires about 3.5 volts of headroom. The extended range will be useful when measuring sensors driven, for example, by the auxiliary amplifier which may go above the 6.2 V potential. In addition, the PNP input stage will continue to operate normally with common-mode voltages of several hundred mV, negative, with respect to common. This feature accommodates self-generating sensors, such as thermocouples, which may produce small negative normal-mode signals as well as common-mode noise on “grounded” signal sources. Figure 9. Functional Flock Diagram AUXILIARY AMPLIFIER The Auxiliary Amplifier is included in the AD693 as a signal conditioning aid. It can be used as an op amp in noninverting applications and has special provisions to provide a controlled current output. Designed with a differential input stage and an unbiased Class A output stage, the amplifier can be resistively loaded to common with the self-contained 100 Ω resistor or with a user supplied resistor. As a functional element, the Auxiliary Amplifier can be used in dynamic bridges and arrangements such as the RTD signal conditioner shown in Figure 17. It can be used to buffer, amplify and combine other signals with the main Signal Amplifier. The Auxiliary Amplifier can also provide other voltages for excitation REV. A –5– 5 Page AD693 Figure 19. Thermocouple Inputs with Cold Junction Compensation Table II. Thermocouple Application—Cold Junction Compensation POLARITY MATERIAL AMBIENT TYPE TEMP RCOMP RZ 30 mV 60 mV TEMP TEMP RANGE RANGE + IRON – CONSTANTAN J 25° 75° 51.7 Ω 53.6 Ω 301K 294K 546°C 1035°C + NICKEL-CHROME 25° 40.2 Ω 392K 721°C — _ NICKEL-ALUMINUM K 75° 42.2 Ω 374K + NICKEL-CHROME 25° 60.4 Ω 261K E 413°C 787°C – COPPER-NICKEL 75° 64.9 Ω 243K + COPPER – COPPER-NICKEL 25° T 75° 40.2 Ω 45.3 Ω 392K USE WITH GAIN >2 340K via a set of thermocouple tables referenced to °C. For example, the output of a properly referenced type J thermocouple is 60 mV when the hot junction is at 1035°C. Table II lists the maximum measurement temperature for several thermocouple types using the preadjusted 30 mV and 60 mV input ranges. More convenient temperature ranges can be selected by deter- mining the full-scale input voltages via standard thermocouple tables and adjusting the AD693 span. For example, suppose only a 300°C span is to be measured with a type K thermo- couple. From a standard table, the thermocouple output is 12.207 mV; since 60 mV at the signal amplifier corresponds to a 16 mA span at the output a gain of 5, or more precisely 60 mV/ 12.207 mV = 4.915 will be needed. Using a 12.207 mV span in the gain resistor formula given in “Adjusting Input Span” yields a value of about 270 Ω as the minimum from P1 to 6.2 V. Adding a 50 Ω potentiometer will allow ample adjustment range. With the connection illustrated, the AD693 will give a full-scale indication with an open thermocouple. ERROR BUDGET ANALYSIS Loop-Powered Operation specifications refer to parameters tested with the AD693 operating as a loop-powered transmitter. The specifications are valid for the preset spans of 30 mV, 60 mV and those spans in between. The section, “Components of Error,” refers to parameters tested on the individual functional blocks, (Signal Amplifier, V/I Converter, Voltage Reference, and Auxiliary Amplifier). These can be used to get an indication of device performance when the AD693 is used in local power mode or when it is adjusted to spans of less than 30 mV. Table III lists the expressions required to calculate the total error. The AD693 is tested with a 250 Ω load, a 24 V loop supply Table III. RTI Contributions to Span and Offset Error RTI Contributions to Offset Error Error Source Expression for RTI Error at Zero IZE Zero Current Error IZE/XS PSRR Power Supply Rejection Ratio (\|VLOOP – 24 V\| + [\|RL – 250 Ω\| × IZ]) × PSRR CMRR Common-Mode Rejection Ratio \|VCM – 3.1 V\| × CMRR IOS Input Offset Current RS × IOS RTI Contributions to Span Error Error Source XSE Transconductance Error XPSRR Transconductance PSRR1 XCMRR Transconductance CMRR XNL Nonlinearity IDIFF Differential Input Current2 Expression for RTI Error at Full Scale VSPAN × XSE \|RL – 250 Ω\| × IS × PSRR \|VCM – 3.1 V\| × VSPAN × XCMRR VSPAN × XNL RS × IDIFF Abbreviations IZ IS RS RL VLOOP VCM VSPAN XS Zero Current (usually 4 mA) Output span (usually 16 mA) Input source impedance Load resistance Loop supply voltage Input common-mode voltage Input span Nominal transconductance in A/V 1The 4–20 mA signal, flowing through the metering resistor, modulates the power supplyvoltage seen by the AD693. The change in voltage causes a power supply rejection error that varies with the output current, thus it appears as a span error. 2The input bias current of the inverting input increases with input signal voltage. The differential input current, IDIFF, equals the inverting input current minus the noninverting input current; see Figure 2. IDIFF, flowing into an input source impedance, will cause an input voltage error that var- ies with signal. If the change in differential input current with input signal is approximated as a linear function, then any error due to source impedance may be approximated as a span error. To calculate IDIFF, refer to Figure 2 and find the value for IDIFF/ + In corresponding to the full-scale input voltage for your application. Multiply by + In max to get IDlFF. Multiply IDIFF by the source impedance to get the input voltage error at full scale. REV. A –11– 11 Page

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