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PDF A4401 Data sheet ( Hoja de datos )
| Número de pieza | A4401 | |
| Descripción | Automotive Low Noise Vacuum Fluorescent Display Power Supply | |
| Fabricantes | Allegro Micro Systems | |
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A4401
Automotive Low Noise Vacuum
Fluorescent Display Power Supply
Features and Benefits
▪ Multiple output regulator
▪ 7 to 40 V input supply
▪ Low EMI conducted and radiated emissions
▪ Adaptive quasi-resonant turn on/off control
▪ Minimal number of external components
▪ Enable input which can be driven with respect to the
battery voltage
Package: 8-pin narrow SOIC (suffix L)
Description
This device provides all the necessary control functions to
provide the power rails for driving a vacuum fluorescent
display (VFD) using minimal external components. The power
supply is based on a quasi-resonant, discontinuous flyback
converter, operating near the critical conduction boundary. A
novel adaptive turn-on control scheme is used to optimize the
turn-on and turn-off phase of the MOSFET, to reduce EMI
emissions while minimizing switching losses.
The converter is self-oscillating, operating at switching
frequencies depending on the input voltage, load, and external
components. An onboard linear regulator that is powered
directly from the battery provides the housekeeping supply,
avoiding the need for complex bias supplies.
Internal diagnostics provide comprehensive protection
against overloads, input undervoltage, and overtemperature
conditions.
The A4401 is supplied in an 8-pin narrow SOIC package
(suffix L), which is lead (Pb) free, with 100% matte-tin
leadframe plating.
Approximate Scale 1:1
Typical Application
+VBAT
ECU
VIN
A4401
LX
GD
ISS
EN
GND
VA
COMP
0V
0V
VFD
A4401-DS
1 page  
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A4401
Automotive Low Noise Vacuum
Fluorescent Display Power Supply
Functional Description
Basic Operation
A peak current-mode control scheme is used to regu-
late one of the converter outputs, which will typically
be the highest output voltage. The regulated output
voltage is potentially divided down and fed into a
Gm stage, where the resulting error signal acts as the
control reference. This reference signal is compared
against the signal that is produced by the inductor
magnetization current flowing through the sense resis-
tor.
As shown in figure 1, at the beginning of a switching
cycle, the external MOSFET, Q1, is turned on. After
the sense resistor signal reaches the control reference
amplitude, the PWM comparator resets the synchronous
rectification (SR) latch and turns off the MOSFET.
When the MOSFET is turned off, the voltage on the
LX node rises until the voltage clamps at the bat-
tery voltage, VBAT , plus the reflected output voltage,
VOUT(RFL). The secondary rectification diodes are
forward biased and the energy stored in the coupled
inductor is released to the output circuits. During this
period, the current through the inductor decreases lin-
+V Coupled inductor goes discontinuous;
resonant ring occurs
VOUT(RFL)
early. As the current falls to 0 A, a resonance is set up
between the primary magnetizing inductance and any
capacitance appearing between the drain and ground.
A damped voltage ringing occurs, which resonates
around the battery voltage, VBAT. As the resonant ring
swings negative, the adaptive turn-on circuit moni-
tors to detect the point at which the voltage reaches a
minimum. At this point the MOSFET is commanded
on, thereby minimizing the turn-on losses. Also, the
relatively slow resonant dV/dt helps to reduce EMI.
In most applications, the converter will be operated
with a battery input voltage of 13.5 V. To optimize
the performance of the regulator at this voltage, the
magnetics can be designed to force 0 V across the
MOSFET at turn-on. This minimizes switching losses
and perhaps more importantly reduces EMI caused by
voltage ringing due to the drain to ground capacitor
resonating with the primary inductance. The voltage
resonance at the MOSFET turn-off can be reduced by
a simple low-loss R-C snubber, as described in the
Electromagnetic Interference section.
If a small enough load is applied to the outputs, and
the output of the Gm stage falls below a certain level,
the converter will enter a burst mode of operation.
Burst mode reduces switching losses while maintain-
ing regulation of the outputs.
VBAT
0
MOSFET
turns off
+I
VOUT(RFL)
MOSFET
turns on
Current released from
coupled inductor into
output circuit
½ resonant
period
Figure 1. External MOSFET voltage and current
Current builds up in
primary winding of
coupled inductor
During startup, assuming the battery voltage is above
the turn-on threshold and the EN input is enabled, the
controller turns on. A soft start circuit controls the ref-
erence voltage, limiting the amount of current drawn
on the input and the amount of charge transferred to
the output, preventing voltage overshoot. During the
initial phase of the soft start, very little or no voltage is
present on the output. This means that there will be no
resonant phase and the converter will operate in con-
tinuous-conduction mode. The converter effectively
operates in constant-current mode until regulation is
achieved.
Allegro MicroSystems, Inc.
5
115 Northeast Cutoff, Box 15036
Worcester, Massachusetts 01615-0036 (508) 853-5000
www.allegromicro.com
5 Page 
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A4401
Automotive Low Noise Vacuum
Fluorescent Display Power Supply
then:
Ig(approx) =
μO × Ae × NP²
LP
.
(27)
Because of flux fringing effects, the above air gap
should be modified, according to the following formu-
las. Given:
F=
Ig(approx)
A½e
×
ln
⎛
⎜⎜
⎝
2×G ⎞
Ig(approx)
⎟⎟
⎠
.
(28)
where lg(approx) is the previously calculated approxi-
mate air gap, and G is the bobbin width (the effective
winding width).
then the total air gap can now be found:
Ig = Ig(approx) × (1 + F) .
(29)
Note that most ferrite core manufacturers provide a
limited number of air gap sizes. It is therefore recom-
mended to select a standard size. Size is indicated
in terms of the Al factor, which is expressed in L/N2
units. The Al factor can be derived from the above two
formulae.
To minimize flux leakage effects, it is recommended
that the air gap should be located on the center limb.
If, however, a distributed air gap is used, the air gap
figure should be divided by two.
Some applications require an AC filament output.
Typically this may be a center tapped winding with
the center tap held at some bias voltage. During the
MOSFET off-time, the output voltage of the control
winding is simply reflected through the turns ratio of
the magnetics.
During the on-time of the MOSFET, the windings are
driven as a forward converter because there is no rec-
tifying diode to isolate this action. The voltage during
this interval is simply the battery voltage transformed
by the turns ratio of the filament winding and the pri-
mary winding. This means that as the battery voltage
varies, there will be a variation in the filament voltage.
However, this variation will be less than those aris-
ing in other converter topologies because, during the
MOSFET off-time, the voltage is regulated.
The voltage amplitude across the filament winding is:
where:
VFIL = VBAT ×
NSF
NP
+ VOUT1 ×
NSF
NS1
,
(30)
NSF is the number of turns on the filament winding,
NS1 is the number of turns on the main controlled
output winding, and
VOUT1 is the output voltage of the main controlled
output.
It is probably desirable to optimize the filament volt-
age at nominal battery conditions; for example, at
VBAT = 13.5 V.
Due to the low voltage out, the number of integer turn
combinations is limited. So, the filament voltage may
not be exact. The turns range may only be 2 or 3.
The magnetic wire sizing for each winding is deter-
mined by the ampere-turns ratio as a proportion of the
total ampere-turns of all the windings. The amount of
bobbin area available for the windings is influenced by
the amount of insulation required, the winding con-
struction technique, and the packing density of the cir-
cular wire. A conservative utilization factor is 0.5, that
is, 50% of the bobbin window area filled with copper.
The rms current of each winding has to be determined.
The worst case condition is at minimum input voltage
and maximum load.
The primary winding current is identical to the current
flowing in the current sense resistor (see the Current
Sense Resistor Selection section). The current in each
of the other output windings can be found as follows.
Given:
IPK =
2 × IOUT
D'(max)
,
(31)
Allegro MicroSystems, Inc.
11
115 Northeast Cutoff, Box 15036
Worcester, Massachusetts 01615-0036 (508) 853-5000
www.allegromicro.com
11 Page | ||
| Páginas | Total 17 Páginas | |
| PDF Descargar | [ Datasheet A4401.PDF ] | |
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